This invention pertains to a digital adaptive equalizer for equalizing signal distortion which varies with the transmission distance and setting gain by a digital signal and a timing controller provided in a transmission interface device. The digital adaptive equalizer is provided, for instance, in a digital subscriber line transmission interface device for performing high-speed data transmission by multi-value pulse signals, e.g. having one of four value levels in the amplitude direction, simultaneously being transmitted in both emitting and receiving directions over existing metallic pair cable telephone subscriber lines.
It has become a wide-spread practice to transmit modulated digital signals over existing analog telephone lines, etc. Here, when high-speed digital transmission is performed, it is essential to equalize the loss-frequency characteristics (hereafter referred simply as loss characteristics) of the signal amplitude in the metallic cable connecting a subscriber terminal with a switching system at an exchange station.
At present, with the advances of digital signal processing techniques and LSI technologies, research is aimed at developing a digital signal processing LSI (DSP) to realize an amplitude equalizer for equalization, including the functions of a balancing network or a variable attenuator.
Although digital filtering is used when a digital signal processing LSI compensates the loss characteristics of a cable, the following points need to be remembered.
Since the distance between a subscriber terminal and its exchange station is not constant, each subscriber has a different cable length, which causes the loss characteristics of the cable to change, as shown in FIG. 1. Accordingly, an amplitude equalizer in a digital subscriber line transmission interface device needs to be an adaptive equalizer capable of changing its frequency characteristics to be able to cope with varying loss characteristics from subscriber terminals to their exchange station. An adaptive equalizer is defined as an equalizer capable of changing its own frequency characteristics by changing the parameters of the functions defining the equalizer itself according to the cable length.
The frequency band generally used is 80 kHz and the transmission band is from 0 to 80 kHz, including a direct current component.
FIG. 2 shows the configuration of a conventional digital subscriber line transmission interface device.
After a hybrid transformer 1 separates the signal transmitted from a subscriber through an analog pair cable or transmission cable 2, a low pass filter (RLF) 3 on the receiving side limits the band of the signal to the frequency bandwidth of not more than 1/2 of the sampling frequency of an A/D converter, for example, 15.36 MHz. The transmission signal produces a pulse at 80 kHz and this spectrum of the transmission signal is condensed to 0 to 40 kHz theoretically. Thus a frequency more than 40 kHz may be cut off by a low pass filter. However, practically speaking, when a frequency more than 40 kHz is completely cut off, the wave form changes greatly due to influence of a delay characteristic of the filter and thus a filter for cutting off a frequency more than 120 kHz is used. Namely, more than 3/2 of the sampling frequency is cut off. Then, an over-sampling type A/D converter A/D comprising a modulator 4 and a decimation filter 5 converts the signal into a digital reception signal and inputs it to a DSP 6. The over-sampling type A/D converter which is used as the A/D converter performs a sampling at a frequency several tens to several hundreds higher than the basic sampling frequency (which is 80 kHz, for example) and converts an input signal to a digital signal of 1 bit, 15.36 MHz, for example. The modulator 4 comprises an integrator, comparator, current delay circuit, and one bit D/A converter. Then, the high frequency noise of the output of the modulator 4 is removed therefrom by a decimation filter 5 and the signal is changed to a low-speed digital signal and is thereby converted to a low-speed digital signal of multiple bits (e.g., 14 bit, 80 kHz). The digital reception signal passes through a subtracter 7, an amplitude equalizer (AEQL) 8 and a decision feedback equalizer (DFE) 9. The amplitude equalizer (AEQL) is aimed at correcting the change of the amplitude of the signals transmitted along a cable.
The frequency loss characteristics of the cable changes depending on the length of the cable. The loss change by 15 dB even in a low frequency band and it is necessary to change an amplitude coefficient of a gain of the equalizer between 1 and 5. The amplitude varies more widely in higher frequency bands. Therefore, the amplitude of the received signal varies widely. The amplitude equalizer (AEQL) 8 corrects such variations.
In contrast, the decision feedback equalizer (DFE) is aimed at waveform-shaping a tail portion of the received waveform (postcursor) by adjusting the amplitude and the frequency characteristics and thus perform a fine correction.
The amplitude equalizer also aids the decision feedback equalizer but the decision feedback equalizer cannot perform the function of the amplitude equalizer. The decision feedback equalizer cannot control the portion of the main amplitude (main cursor). After the digital reception signals pass through the decision feedback equalizer (DFE) 9, the DSP 6 outputs the digital signal as digital reception signal 10.
On the other hand, having undergone necessary digital signal processings (such as binary-to-multiple value conversions) by a coder COD 11, the digital signal supplied to the DSP 6 is outputted as digital transmission signal 12. Thus, a D/A converter 17 makes the digital transmission signal outputted from the DSP 6 analog, and a low pass filter (SLF) 13 on the receiving side limits less than about 3/2 of the sampling frequency. After going through a driver circuit (DRV) 14 and the hybrid transformer 1, the analog transmission signal is sent from the analog pair cable 2 to subscribers.
Here, since a part of the transmission signal bound for the analog pair cable 2 from the hybrid transformer 1 affects the receiving side as an echo component and is inputted, as a part of the digital reception signal, to the DSP 6, the receiving side needs to cancel the above echo component. Hence, an echo canceler (EC) 15 generates, as an echo replica component, the above component from the digital transmission signal, which the subtracter 7 subtracts from the digitized reception signal, so that the echo component is canceled. In this case, the decision feedback equalizer (DFE) 9 is connected to the output of the amplitude equalizer (AEQL) 8 on the receiving side and outputs the error between the reception signal and a decision symbol value which is required for generating the echo replica component in echo canceler 15. The error signal from the decision feedback equalizer a is input to the echo canceler 15. The error signal is provided by the symbol values (.+-.1, .+-.3) corresponding to the difference between the output value and the input value of the decision feedback equalizer 9 the polarity alone may be used. The decision feedback equalizer (DFE) 2 also has a function of adaptively controlling parameters of a transversal filter in the equalizer 8 to remove intersymbol interference of the signals received from the sending station. For this purpose an error between the received signal and the symbol value of the decision result is obtained.
Combinations of pieces of the hardware of the DSP 6 and their controlling microprograms realize the respective functions of the above DSP 6, or hardware can be used to realize these functions without using microprograms.
The amplitude equalizer (AEQL) 8 in the DSP 6 equalizes (adjusts) the loss of the digital reception signals for loss having frequency characteristics in the analog pair cable 1. Further, the amplitude equalizer (AEQL) 8 ordinarily has an AGC (automatic gain control) function. As shown in FIG. 1, the the loss frequency characteristics of the cable increase as the length of the cable increase. Since the loss in low frequencies becomes commensurately larger, not only the gradient but also the gain needs to be increased, which is performed by the AGC function.
A conventional amplitude equalizer 8 uses selected one of three to five types of filtering coefficients predesignated for the cable length of the analog pair cable 2, so that the amplitude equalizer 8 has equalization characteristics (frequency transmission characteristics) in accordance with the cable length, such as those shown in FIG. 3, corresponding to the loss frequency characteristics shown in FIG. 1.
At this time, per a conventional method, a sum-of-squares calculator 16 shown in FIG. 2 obtains the electric power of the signals inputted to the amplitude equalizer 8 for every predetermined time period by calculating the sum of the squared amplitude of the signals, so that the proper filtering coefficient, i.e. the equalization characteristics, is obtained commensurately with the value of the electric power. For example, time periods during which a sum-of-squares is small result in filtering coefficients having steep high pass characteristics and large gains in low frequencies corresponding to long distance cables, because the signal amplitudes for such time periods are generally smaller.
However, since the above described prior art example of the amplitude equalizer 8 selectively uses several kinds of discrete filtering coefficients, it cannot precisely adjust the filtering coefficients in response to continuous small changes in the cable lengths. That is, a problem remains that a significant error exists in the digital reception signal after equalization, when the necessary equalization characteristics are somewhere in the middle of a pair of predesignated equalization characteristics and do not exactly match any of them. For example, when filters whose low frequency gains changes from 0 to 16 dB at an interval of 2 dB are provided as a plurality of filters, an error of 1 dB remains in a case of a cable with 7 dB loss.
A further problem is that the calculation of the sum-of-squares is necessary and thus the load of DSP 6 increases.
An even bigger problem is that calculation of the sum of the squares needs to be repeated several times until the phases match, because the obtained amplitudes are not exact when the sampling phase is not matched with the signal phase.
In response to the above problems, a digital variable filter which calculates filtering coefficients by having a special conversion function converting a parameter for the cable length has been proposed, but it is not applicable to a transversal filter, because the conversion function is so special. Further, although this applicant disclosed a digital adaptive equalizer applicable to a transversal filter in the 1990 Japanese Patent Application No. 53787, since that invention was premised on the decibel indication of the loss characteristics proportional to the cable length, the invention has a problem that it cannot be applied to other loss characteristics.
Another prior art device obtains an average value of amplitudes at a plurality of times which do not overlap each other and compares it with a reference level. This is performed at two stages comprising a coarse adjustment for changing a level exponentially and a fine adjustment for changing a level at a fine step. This prior art device requires a serial type multiplier for performing a multiplication of 16 bits.times.8 bits and thus has to provide an exclusive hardware.
A decision feedback equalizer equalizing intersymbol interference on the time axis is used for a digital signal receiving device. Also, a configuration is adopted such that an automatic gain control amplifier or a .sqroot.f equalizer compensates changes in attenuation characteristics or frequency characteristics of a transmission path. It is desired to realize such a configuration economically.
A conventional line equalizer has a configuration such as that shown in FIG. 4, for example. An A/D converter 21 converts a reception signal having waveforms corresponding to transmission codes e.g. to a 10-bit digital signal, which an automatic gain control amplifier 22 amplifies to a predetermined level, based on operations for reception signal power. A .sqroot.f equalizer (EQL) 23 equalizes the attenuation characteristics of the transmission paths, and a slicer 24 providing a decision threshold eliminates intersymbol interferences.
An adder 25, a slicer 24 and an equalizing part (DFE) 26 form the decision feedback equalizer 27. The equalizing part 26 generates intersymbol interference components to be supplied to the adder 25, based on the decisions made by the slicer 24.
FIG. 5 is a block diagram of a conventional decision feedback equalizer 27, described earlier, having an n-tap configuration, where 28 denotes an input terminal, 29 denotes an output terminal, 30 denotes an adder, 31 denotes a slicer (DEC), 32 denotes an adder, 33 denotes a tap coefficient updater, 34 denotes an adder, 35-1 through 35-n denote lag elements (T), 36-1 through 36-n denote coefficient multipliers.
A case in which AMI codes are used as transmission path codes is explained as an example. An adder 30 adds the reception signal X.sub.k at time k to the intersymbol interference component R.sub.k having the negative sign to produce the equalized signal F.sub.k, and the slicer 37 determines whether the reception signal is ".+-.1" or ".+-.3" through level decisions. Lag elements 35-1 through 35-n sequentially lags the decision output signal a.sub.k by one baud rate period. Output signals from respective lag elements are supplied to coefficient multipliers 36-1 through 36-n as well as tap coefficient updater 33.
The tap coefficient updater 33 controls the coefficients of the respective coefficient multipliers 36-1 through 36-n, so that error signals e.sub.k are minimized. The respective coefficient multipliers 36-1 through 36-n multiply the tap coefficients C.sub.1k through C.sub.nk by the outputs from the lag elements 35-1 through 35-n, which products are added by the adder 34. Then, the adder 30 adds the intersymbol interference components R.sub.k of the result of the addition, so that the intersymbol interference components R.sub.k are subtracted from the reception signals X.sub.k, thereby eliminating the intersymbol interference included in the reception signal X.sub.k.
FIG. 6 illustrates the relations between a reception signal single pulse response and tap coefficients. C.sub.1 through C.sub.5 are intersymbol interference components for single pulse responses. By forming tap coefficients C.sub.1k through C.sub.5k which are the same as those components, and by having the adder 34 generate intersymbol interference components R.sub.k, the adder 30 can eliminate intersymbol interference. The automatic gain control amplifier 22 in the conventional line equalizer described earlier amplifies the reception signal to a predetermined level based on power detection of the reception signals. A .sqroot.f equalizer (EQL) performs equalizaton through peak detection, for example. Hence, its configuration is more complex than those of the amplifiers having fixed gain or equalizers having fixed characteristics.
As described above, it has become wide spread practice to transmit digital signals modulated for transmission over analog telephone lines. This requires a digital subscriber line transmission interface unit for restoring the original signal waveforms distorted by the transmission characteristics of metallic cables.
A digital subscriber line transmission interface unit often has a function of simultaneously transmitting and receiving digital data quantized to four values (e.g. .+-.1 and .+-.3) in amplitude at a transmission speed of 80 kbaud (kilo bauds). There are two types for this kind of device, one being a network side device (Line Terminator: hereafter abbreviated as LT) and another being a terminal side device (Node Terminator: hereafter abbreviated as NT).
It is crucial to synchronize the timings of the actions over the entire network for enabling signal transmission. In this case, the master clock on the network side becomes the reference. The NT receives the signals emitted from the LT based on the reference, and the NT's timing controlling circuit acts based upon the received signals, so that the timings on the NT's side are set. Since the NT sends signals at the timings so determined, the frequencies at the timings when the LT receives signals match the frequencies of the timings when the NT sends signals, but the phases are different. The phase difference is determined by the lag time resulting from the cable length and the difference between the reception timing and the emission timing at NT. Thus, the LT needs to set the phase difference to the optimal value by adjusting it when communication begins.
Meanwhile, a digital subscriber line transmission interface unit needs to set coefficients of the transversal filter in a decision feedback equalizer and an echo canceler, as well as the NT's timing adjustments. Although almost all of these coefficients are configured to be able to change adaptively, they need to be set initially, for which generally the LT and the NT mutually send training signals to each other for a certain period of time, thereby receiving each others' training signals, which sets timings and filter coefficients.
These adjustments are divided into the adjustment of the echo canceler and the adjustment of the receiver circuit, such as the adjustment of the reception timings and the decision feedback equalizer. That is, the emission training pulses of the near end must adjust the coefficients of the echo cancelers, whereas the training pulses of the far end must adjust the reception system circuit.
Since there are cases in which the reception system circuit and the echo canceler cannot be simultaneously adjusted in the initial training stage, in reality, the NT sends training pulses, after the NT ceases outputting training pulses, the LT outputs training pulses, and then both output training pulses simultaneously.
In this case, the NT uses the training pulses it outputs by itself and adjusts the echo canceler at its own clock timings. Although the LT can receive the training pulses from the NT at this time, since it is not defined to make the pulse numbers sufficiently large, the LT's reception system cannot be adjusted during this period.
Next, the LT adjusts the echo canceler at clock timings of its own, and the NT adjusts the reception system, when only the LT outputs training pulses.
Subsequently, the NT outputs training pulses, while the LT keeps outputting training pulses. The LT's reception system circuit is adjusted during this time period. The NT's reception system circuit has already adjusted the timings at which the NT outputs training pulses per the training pulses from the LT, so that the timings match the correct frequencies, i.e. the network frequencies. Hence, the timing adjustments for the LT's reception system circuit are nothing but matching the phases.
Since the NT at the terminal end needs to change the clock frequencies of its own in accordance with the timings for the received pulses, the emission frequencies change accordingly. However, since the time difference between the emission timing and the reception timing can be set arbitrarily, the echo canceler need not be adjusted any more by setting the emission timing so that the difference becomes the same as that when the NT's echo canceler is adjusted.
However, since the LT has set its clock timing from the network, it cannot change its emission timings in correspondence with the received training pulses. But instead, either the echo canceler already adjusted needs to be readjusted per the timings of the received training pulses or the sampling signals for the echo canceler after the echo is canceled need to obtain the signals at the timings when their phases are changed.
The following is a description of a such conventional example of a timing controlling device.
After the LT trains the echo canceler with the training pulses sent from the LT, the signals are received for training at the timings when the echo canceler performs training. Thereafter, the timing is gradually changed so that the error is minimized. By gradually changing the timings, the tap coefficients for the echo canceler change in connection with the timing changes, the echo canceler can maintain the trained state. When the echo canceler maintains the trained state, since other circuits such as the decision feedback equalizer can be optimized relatively easily, when the timings do change, circuits other than the echo canceler such as the decision feedback equalizer immediately follows the new timings.
As so far described, since the respective circuits in the reception system can observe the output errors e.g. from a decision feedback equalizer, the timings at which the errors are minimized become the best reception timings.
The following is a description of another conventional example of the timing control device.
After the echo canceler is trained, by supplying the received signals through the echo canceler and then through lagged filters having fixed lag periods, the timings are changed. In this case, the combination of constant delay time filters minimizes the error obtained, by changing the connections among a plurality of lag filters having different fixed lag periods.
Here, the relationships between the timings and the lagged periods are briefly explained by referring to FIGS. 7A, 7B and 7C. FIGS. 7A to 7C show an example of digital input signal series, which show single pulse response waveforms. Since the values actually obtained are sampling values, only the values at the bold lines at the timings shown by upward pointing arrows in FIG. 7C are obtained. Since the input signal column shown in FIG. 7A has large amplitudes at a plurality of timings, the intersymbol interference cannot be set to zero. A digital filter can lag the input signal series per its phase delay characteristics. The phase delay characteristics of the digital filter of the minimum phase shifting type are automatically set when the amplitude characteristics are determined. Thus, a mere use of the minimum phase shifting type filter cannot cut, for example, only a high frequency component without causing distortion. Thus, a filter having a transfer function including a term for distortion is generally used for shifting the times, as shown in FIGS. 7A and 7B, without changing the waveforms of input signals. FIG. 7B shows an output signal series obtained by a variable lag filter causing the input signal series shown in FIG. 7A to lag by the time equivalent to a half of the sampling cycle (the cycle period of the decision timing shown in FIG. 7C). Since the output amplitudes at times other than predetermined timings corresponding to FIG. 7A are almost zero, the timing adjustments are known to have worked well.
As described earlier, a plurality of lag filters respectively having average and different phase delay characteristics are provided so that, when their connections are changed to find the combination that minimize error, the timing adjustments are completed. Since the echo canceler need not be readjusted in this case, the time required for timing adjustments is shortened.
Of the conventional methods for timing adjustments, the first prior art device described previously requires the entire reception system circuit to be adjusted by readjusting the echo canceler, since the optimal timings for the reception system circuits are not known when the echo canceler is adjusted.
However, there are cases in which timings are greatly different in the beginning. The initial adjustments of the LT takes a significantly long time, and the hardware size increases because of a need for storing a large scale program.
On the other hand, although the second prior art device can cause the time required for timing adjustments to be shortened and the processing programs can be simplified, lag filters having fixed lag times, such as 1/2, 1/4, 1/8, 1/16, 1/32, . . . of the sampling period need to be provided, which causes an increase in filter size. When these processes are performed by digital signal processings, they don't directly cause an increase in hardware size. However, the degree of the entire filters becomes higher, and the processing volume for the operation necessarily increases, which in turn causes a problem in that as the hardware increases so does the power consumption for executing the process.